Complementary pair antenna element groups



June 10, 1969 r K. e. SCHROEDER 3,449,751

COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20, 1965 Sheet of 10 COMPLEMENTARY ELEMENT +1'X FIG l6 REGULAR ELEMENT 25a 2 2 25a 4 l u Z|-IZ||L z -z 2| 12'' L INSERTION L033 [)8 miruo KLAUS G. SCHROEDER o 10 I5 2b as 30 35 FREQUENCY MC F G I? o 5 IO I5 so FIG '8 FREQUENCY MC INVENTOR.

/ ATTORNE s June 10, '1969 K. s. SCHROEDER COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 2.0. 1965 Sheet FIG 3 INVENTOR. KLAUS 'G. SCHROEDER ATTORNEYS" June 10, 1969 K. G. SCHROEDER 3,449,751

COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20, 1965 Sheet 3 0! 1o FIG 5 0% zoo INVENTOR. KLAUS G. SCHROEDER f ATTORNE S June 10, 1969 K. e. SCH ROEDER 3,449,751

COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20. 1965 Sheet 4 01 1o ww f k ATTORNE S June 1969 K. e. SCHROEDER 3,449,751

COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20. 1965 Sheet :5 of 10 FIG l0 200 250 300 350 400 450- FREQUENCY Mc INVEN'IOR. KLAUS- .6. SCHROEDER ATTORN Y? June 1969 K. G. SCHROEDER 3,

COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20, 1965 Sheet 6 of 10 I=COMPLEX REFLECTION COEFFICIENT (I=u+jb|1"| l) 1 FIG l5 INVEN'IOR. KLAIJS e. SCHROEDER K ATTORNE s June 10, 1969 K. e. SCHROEDER Sheet ATTORNEY" June 10, 1969 K. a. SCHROEDER 3,

' COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20, 1965 Sheet 8 of 10 --FEED BEAM DIRECTION INVEN'I'OR. KLAUS G. SCHROEDER AT TORNEYS June 10, 1969 K. G. SCHROEDER 3,449,751

COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20, 1965 Sheet 9 of 10 2 my mm @Q FIG 26 INVEN'IOR. KLAUS 6. SCHROEDER I {1' AT TORNE' YS June 10, 1969 K. G. SCHROEDER 3,

COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Filed Sept. 20, 1965 Sheet /0 Of 10 INVEN'IOR.

KLAUS G. SCHROEDER AT TORNEVYS United States Patent 3,449,751 COMPLEMENTARY PAIR ANTENNA ELEMENT GROUPS Klaus G. Schroeder, Dallas, Tex., assignor to Collins Radio Company, Cedar Rapids, Iowa, a corporation of Iowa Filed Sept. 20, 1965, Ser. No. 488,541 Int. Cl. H01q 21/00, 1/50, 19/12 US. Cl. 343-727 20 Claims ABSTRACT OF THE DISCLOSURE A new class of broadband antennas with automatic impedance matching through combining two elements as a complementary pair in such a way that the reflection coeflicients cancel. This is accomplished by using two antenna elements with complementary imepance characteristics and feeding them with a 180 degree hybrid so that the radiation produced by the two elements is additive in the desired propagation direction.

This invention relates in general to various antenna systems with broadband operational capabilities, and in particular to antenna systems utilizing complementary paired element groups with each pair fed by a 180 degree hybrid and with each element of respective pairs of elements radiation mutually coupled to a relatively high degree.

With large, in terms of wavelength, unidirectional antenna arrays in the HF, VHF and UHF frenquency ranges where more than approximately an octave of bandwidth is required, the untuned element VSWR (voltage standing wave ratio) in front of a reflector is usually around 2:1. This results, with many existing antenna systems, in problems with large amounts of reflected power disturbing, for example, phasing circuit mechanisms (particularly under beam steering conditions) which can also overload a transmitter. With applicants complementary pair element groups reflected energy is channeled into assigned terminations providing a well matched impedance for circuit element pairs excitable as effectively one element in a high gain broadside array. Such structure provides for ultimate limitation in operational bandwidth as determined by the element pattern and with a desired expansion approximately to the 4:1 frequency range.

A new class of broadband antennas is provided which automatic impedance matching is achieved by combining two elements in such a way that the reflection coefficients cancel. Applicant has accomplished this by selecting two antenna elements with complementary impedance characteristics and feeding them with a 180 degree hybrid so that the radiation produced by these two elements is additive in the desired direction and with such a pair of elements called a complementary pair element group. Various antenna system complementary pair element groups providing such operation in appropriately designed relation are a vertical dipole and a folded dipole complementary pair, and an endfire monopole complementary pair structure, a structure that is a near optimum radiator for vertically polarized steerable antenna arrays, such as super power high frequency transmitting arrays. Such complementary pair element groups are quite useful in many other antenna systems including, for example, the linear and circularly polarized feeds, paraboloids and paracylinders, endfire element groups in vertically polarized circular antenna systems, and stacked element groups in endfire arrays with non-uniform element spacing. Among excellent operational applications are communications, high-quality broadcasting, air traffic control, radar, track, spectrum surveillance and radio astronomy. It should be realized that these groups are very useful both in 3,449,751 Patented June 10, 1969 antenna transmitting systems and receiving systems and provide a more economical, highly efficient and much more flexible antenna operational installation than heretofore provided by other existing antenna radiating and receiving systems.

It is, therefore, a principal object of this invention'to provide improved antenna systems with a desired radiation pattern and broadband impedance matching with both substantially independent of installation environmental variances.

Another object is to provide complementary pair element group antenna structures with greatly expanded broadband full range adaptable impedance matching.

A further object is to provide such transmitting and receiving antennas having almost unlimited gain.

Features of this invention useful in accomplishing the above objects include in the various embodiments a degree hybrid feeding the two elements of each pair of elements forming a complementary pair element group in each antenna system. The two elements of each complementary pair have input impedances that are substantially complementary, that is, with the impedance of one being Z /VSWR and the other being Z times VSWR with Z being the characteristic impedance of each respective element as resistances into the respective elements. This results in the impedances being located 180 degrees apart or substantially so on the same VSWR circle of a Smith Chart (such as referred to hereinafter). The two element of each complementary pair are so spaced, positioned and designed as to have a relatively high degree of radiation mutual coupling. There are basically two different types of complementary pairs that may be employed in applicants improved antenna systems including, first, those which are truly complementary in the Babinet sense, such as a slot and dipole, and also those which can be designed to be complementary, such as dipoles and folded dipoles; and secondly, those which are externally complementarized by special circuit techniques.

Specific embodiments representing what is presently regarded as the best mode of carrying out the invention are illustrated in the accompanying drawings.

In the drawings:

FIGURE 1 represents the schematic circuit diagram of a 180 degree hybrid circuit such as used in feeding the various complementary pair element group equipped embodiments;

FIGURE 2, a combination block and schematic illustration of the general impedance relations involved in use of 180 degree hybrids;

FIGURE 3, a perspective view of an endfire complementary pair element group mounted above and fed through a ground plane and equipped with a reflector;

FIGURE 4, a showing of a 180 degree hybrid feeding two 25-ohm terminating effective resistances or characteristic impedances of two of the elements of an element group with the elements made to appear complementary through the use of two coaxial lines with one a quarter of a wavelength of the operational bandwidth center frequency longer than the other;

FIGURE 5, a graph of voltage standing wave ratio (VSWR), of an end-fire monopole complementary pair element group with reflector as shown in FIGURE 3 throu h the operational bandwidth range;

FIGURE 6, a schematic showing of a broadband endfire complementary pair feed circuit;

FIGURES 7, 8 and 9, H-plane pattern graphs at frequencies of 180 mc., 400 mc., and 600 mc., respectively, of an endfire monopole complementary pair element group embodiment with back lobe limitation determined by the reflector size;

FIGURE 10, a percentage power absorbed by difference port load for an endfire monopole complementary pair element group through the frequency range of operation graph;

FIGURE 11, a Smith Chart showing optimized locus of an endfire monopole complementary pair element group front element having an impedance Z equal to 50 ohms;

FIGURE 12, the corresponding plot for the back element through the same frequency range;

FIGURE 13, a plot of resistive loads on a Smith Chart with respect to zero reactant complementary loads;

FIGURE 14, a showing of arbitrary complementary loads with approximately 2:1 and 3:1 standing wave ratios;

FIGURE 15, a Smith Chart overlay on the complex reflection coefficient plane;

FIGURE 16, a plot of the beam impedance characteristics of dipoles and folded dipoles;

FIGURE 17, a plot of measured insertion losses for a high power, high frequency 180 degree hybrid through an extended operational frequency bandwidth range;

FIGURE 18, a graph of measured sum port VSWR with a high power, high frequency 180 degree hybrid through a corresponding frequency range;

FIGURE 19, a perspective view of a vertical dipole and folded dipole complementary pair element group mounted on a reflector and with a ground plane;

FIGURE 20, a broadside reflector antenna array with vertical monopole and folded dipole complementary pair element groups;

FIGURE 21, a broadside reflector array with horizontal dipole and folded dipole complementary pair element groups;

FIGURE 22, a non-uniformly spaced endfire antenna array of vertically stacked folded monopole and dipole complementary pair element groups;

FIGURE 23, a vertical paracylinder antenna with endfire monopole complementary pair element group feed;

FIGURE 24, a paraboloidal reflector antenna using a cross polarized endfire dipole complementary pair element group feed;

FIGURE 25, a fixed bearing circular antenna array of endfire monopole complementary pair element groups with individual element group post-type reflectors;

FIGURE 26, an endfire monopole complementary pair element group equipped Wullenweber antenna;

FIGURE 27, a circular antenna array of endfire monopole complementary pair element groups;

FIGURE 28, a perspective view of a 32-element monopole antenna array with the elements arranged in 16 endfire complementary pair element groups and with a reflector; and,

FIGURE 29, a perspective view of a self-supporting, cylinder miniature slot and dipole complementary pair element group antenna for circularly-polarized UHF transmission.

Referring to the drawings:

With the new complementary pair element groups a broadband impedance match is accomplished by feeding two antenna elements with complementary characteristics from one hybrid junction. This new useful system is embodied in many various antenna environmental embodiments such as element groups including a vertical dipole and folded dipole, and the endfire monopole complementary pair which is a near-optimum radiator for vertically polarized steerable arrays such as super power HF transmitting arrays. These two embodiments are described in detail hereinafter along with various other embodiments to lesser or the same extent. Such embodiments advantageously provide for very broadband operation of high gain broadside reflector antenna arrays and is also extremely useful in a considerable variety of other antenna systems. Examples are linearly and circularly polarized feeds for paraboloids and paracylinders, endfire element groups in vertically polarized circular arrays, and

stacked element groups in endfire arrays with non-uniform element spacing.

Elements in naturally complementary pairs in an antenna system have to be arrayed side by side in back element and forward element planes perpendicular to the jdesired beam direction to provide broadside complementary pair element groups. The following table is a partial summary of complementary element pairs either naturally complementary or with the potential of being so designed and arranged as to be substantially complementary.

TABLE 1 (1) DipoleSlot (2) Vertical dipoleFolded vertical dipole (3) Horizontal dipoleFolded horizontal dipole (4) Vertical monopole-Folded vertical dipole (5) Vertical monopoleFolded horizontal dipole (6) Vertical dipoleFolded vertical monopole (7) Vertical dipoleFolded horizontal dipole (8) Horizontal dipoleFolded vertical monopole (9) Slot-Folded slot (10) Folded dipoleFolded slot (11 DipoleOrthogonal dipole (12) Folded dipole-Orthogonal folded dipole (13) SlotParallel folded dipole (14) Folded slotParallel dipole The impedance characteristics of the elements within each pair do not necessarily rotate about the same point on a Smith Chart and only the vertical monopole locus can be made to rotate around 50 ohms. However, the center points of the impedance characteristics can be transformed to 50 ohms, or any other characteristic impedance. As an example, a narrow slot and thin dipole are complementary with respect to half the intrinsic impedance of free space, or approximately 601- ohms. A complementary pair element group of these elements has to be fed by a hybrid having a characteristic impedance of substantially 601r ohms. Since this is diflicult to achieve, a practical approach is to get down to a 50 ohm line impedance by means of a step-down transformer with approximately a 4:1 transformation ratio. A crucial part of the various complementary pair element group embodiments is a degree hybrid, such as shown in FIGURE 1, various embodiments of which are fairly well known. Such a hybrid, also known to microwave people as the Magic T, has the property that any signal applied to the summation terminals of sum port Z is distributed into two phase components at the terminal pairs Z and Z;*. A signal applied to the Z pair of terminals results in two components for Z and Z which, while they are of equal amplitude, are 180 degrees out of phase. An important property of such hybrids is their ability to average reflection coeflicients. From the emittance matrix of this structure, the sum port Z emittance is usually converted in a conventional way into the sum port impedance.

Letting N =N *=1 and N =n 2 the sum port impedance simplifies to pedances are complementary with respect to the characteristic impedances of the hybrid, i.e.,

21* =Z02/Z1 then the sum port impedance is finally:

and

Equation (3) These impedance relationships are illustrated in FIG- URE 2 with a load impedance Z at one output port and Z /Z at the other output port and with the sum port Z impedance equal to Z The useful frequency range of complementary pair element groups in an antenna environment is limited by the E-plane pattern of the individual elements to approximately 4:1, but this is about the maximum bandwidth of broadside phased arrays. This is because beam steering to say as much as to an angle of 40 degrees requires that element spacing be less than 0.67\ (wavelength) if grating lobes are to be avoided. Since spacing S between the elements has to be substantially For a 4:1 frequency range, the spacing between elements of a complementary pair is then approximately 0.15 wavelength at the lowest frequency, a spacing that should be the minimum tolerable spacing with respect to mutual coupling variation. The fact that colinear complementary pair elements cannot be spaced that closely leads naturally to the selection of parallel radiators. Vertical monopoles above a conducting ground plane are very economical because they do not require balancing networks. They can be fed directly from a coaxial feed system, and further, they have resistances below 50 ohms in the vicinity of first resonance. Thus, no additional transformation is necessary to get the center of the monopole impedance locus approximately at 50 ohms.

An endfire pair broadband monopole embodiment 30 of FIGURE 3 reflector 31 assures high front-to-back ratio at all frequencies but it should be noted that this endfire complementary pair element group can be designed in an antenna system without the reflector. The rear element 32 has a top loading extended planar end 33 for fine adjustment of the sum port Z impedance of the total complementary pair element group circuit network. The impedance locus of the rear element 32 shows a larger VSWR than the front element 34 since the back element is closer to the reflector 31. These optimized elements are now complementarized as is shown in FIG- URE 4, assuming that both element impedances are approximately 25 ohms with the rear element 32 and the forward element 34 being fed through coaxial lines 35 and 36 respectively from a 180 degree hybrid 37. The connections coaxial lines 35 and 36 are with outer sheaths to ground plane 38 and the inner conductors directly to the bottom conical apexes of the feed ends 39 and 40 of the rear element 32 and forward element 34, respectively. With both the rear and forward impedances approximately 25 ohms one of the monopoles can be made to be complementary by feeding it through a coaxial transmission line which is a quarter of a wavelength longer at midband of the frequency range than the length of the line feeding the other element. This is accomplished by making line 36 a quarter of a wavelength longer than the coaxial line 35 in the embodiment 30 of FIGURE 3. Then the 180 degree hybrid 3'7 automatically matches these two impedances together, because the conditions required for the formula Z *=Z /Z are satisfied. Even though this condition is exactly true only at the center frequency, a 3 :1 bandwidth is easily realized, i.e., the complementarization is effective for a feed line differential length from The sum port VSWR of the total endfire monopole complementary pair element group with reflector is shown in FIGURE 5. The solid line represents the sum port VSWR for 50 ohm coaxial terminations at the two output ports. Normally is should not be possible to get a better match than this hybrid VSWR, but due to apparent reactance compensation at 650 me. (which was beyond the hybrid design band) a slightly better VSWR resulted for the complementary pair element group. It is interesting to note that equal-ripple characteristic of the input standing wave ratio, which is due to the various factors contributing to the over-all complementary pair element group performance. At 400 mc., the center frequency of the design band, the standing wave ratio has the expected minimum, because the feed line difierential is exactly M4, and rotates one impedance into a complementary position with respect to the other. At lower frequencies lower minimum VSWRs can be expected, corresponding to the larger bandwidth and better match of the hybrids. FIGURE 6 shows the total near-field beam forming mechanism, which is applicable to both monopole and dipole endfire complementary pair element groups, even though the selfand mutual impedances are different. For broadband patterns a time delay differential 1- is used in the front element feed line, where r is the wave delay between the front and the back elements.

Representative H-plane patterns are shown in FIG- URES 7, 8 and 9, the blacklobe being determined by the limited reflector size. The changes in the H-plane patterns are due to the fact that only at the center frequency f (400 mc.) the time delay T is equal to 4, so that a high front-to-back ratio results in the cardioid pattern. At 0.5 and 1.5 (200 and 600 me.) the time delay differential is equal to g and 3% respectively, the backlobe peak is only down about 3 db from the forward beam peak, resulting in a narrower pattern at 200 mc., and a dip in the center at 600 mc., according to the spacing of the phase center from the reflector. The H-plane patterns can be adjusted by moving the group closer to the reflector, without affecting the group impedance match very much. In FIGURE 10 the power absorbed by the difference port load of the complementary pair element group hybrid is shown as a function of frequency. A maximum matching loss of -20% occurs as expected at the center frequency, because here the self-impedances are maximally complementary, and the mutual impedance (called Z in FIGURE 6) is already smal enough to have little effect on the sum port impedance. At 200 mc., however, very little energy is lost in the difference port. Nevertheless, a good impedance match results at 200 me. of about 116:1 VSWR, as compared to the individual monopole mismatches of about 6:1 VSWR. This, mutual impedance Z has been successfully employed in the over-all input impedance of the complementary pair element group, with minimum matching loss, and with paired element spacing less than 0.7)..

The FIGURE 11 shows an optimized locus of an endfire monopole front element having an impedance Z, equal to 50 ohms in a complementary pair element group embodiment such as the embodiment 30 of FIGURE 3 in an antenna system designed for an operational bandwidth range approximately 200 to 600 megacycles. The impedance locus for the rear element 32 is given in FIGURE 12 and shows a larger VSWR because the back element is closer to the reflector 31. Dimensions of a Working FIGURE 3 embodiment 30 endfire complementary pair element group antenna system include elements 32 and 34 body diameters of 3 inches, bottom conical apex 39 and 40, heights of 2.6 inches, rear element 32 spacing of 8.8 inches from the reflector 31, and forward element 34 spacing an additional 7.5 inches forward from the rear element 32, and with the element 32 top loading extended planar end 33 being 6 inches in diameter.

Referring again to the hybrids of FIGURES 1, 2, 4 and 6, in complementary pair element group antenna system usage, FIGURE 13 is illustrative of resistive loads on a Smith Chart with respect to zero-reactance complementary loads. Load Z is equal to 100 ohms and its complement 2 is 25 ohms. Together, if terminating a 180 degree hybrid, they result in a matched sum port, even though both have a 2:1 VSWR with respect to 50 ohms. Also, Z and Z Z and 2 and finally Z and Zf (an open circuit and a short) result in a matched sum port. In the last example, all the power sent into the sum port gets absorbed in the difference port, because the totally reflected output signals are 180 degrees out of phase. Those examples were all on the real axis. The next example, in FIGURE 14, shows arbitrary complementary loads, with 2:1 and 3:1 standing wave ratios. It can be readily shown that for any pair of output impedances, even if not complementary, the sum port impedance is always the center of a straight line connecting the two impedances on a Smith Chart.

The 180 degree hybrid of FIGURE 1, known in microwave theory as the Magic T, operates in a well-known fashion such that any signal applied to the sum port Z terminals divides into two in-phase components of equal amplitude at the respective Z and Z terminals whereas a signal applied to the Z terminals results in two components at the Z and 2 terminals of equal amplitude but 180 degrees out of phase. A less publicized performance characteristic of such 180 degree hybrids is that the reflection coeflicient T at the sum port Z is equal to the arithmetic mean of the reflection coeflicients, I; and If at the Z and Z terminals, respectively (i.e.,

Hence, the sum port impedance Z appears on the center of a straight line connecting the two output impedances, Z and Z on a Smith Chart. Therefore, if Z and Z are complementary with respect to the system characteristic impedance, Z (i.e., Z =Z /Z then Z is equal to Z Applicant has applied this principle successfully in the development of a matched antenna element group for use in phased arrays. However, it should be noted that a significant improvement in sum port impedance match can also be realized even if the impedances Z and Zf are only partially complementary, a fact easily verified by considering the FIGURE Smith Chart plot. For Z equal to 0.2Z (VSWR of 5.021) and Zf" equal to 22 (VSWR=2.0:1), the sum port impedance Z is 0.7Z (VSWR=1.4:1).

One may prove mathematically the validity of the above-described averaging principle by obtaining from circuit considerations the expression for the sum port impedance, Z in terms of the two output impedances, Z and Zf and comparing this with the expression for the impedance at the center of a straight line connecting Z and Z on the Smith Chart as follows:

From the admittance matrix, the sum port admittance Y;, is shown to be;

Next, the turn ratios will be selected as follows:

Now if the difference port is terminated in the characteristic impedance of the transmission system in which the hybrid is used, (i.e., Z =Z and Equation 1A is normalized with respect to Y (Y =1/Z we obtain the equation for the sum port impedance of a matched hybrid, i.e.,

From Equation 2A, it is seen that the hybrid sum port is perfectly matched to the input transmission line when plex reflection coefficient plane. Consider the arbitrary reflection coetficients F1 and P1 If we define I as the reflection coefficient at the midpoint of a straight line connecting The normalized impedance at this point is given by Equation (3A) If we write I and 11* in terms of 2 and z;* respectively and substitute in Equation 3A we obtain,

1 M iii 1+/2 z1+1 z.*+1 T Pac I Equation (4A) S1mplification of Equation 4A yields By comparing Equations 2A and 5A it is seen that z; =z Hence the hybrid sum port impedance does, in fact, fall on the geometric center of the straight line connecting the two output impedances on the Smith Chart.

A 180 degree hybrid such as the hybrid 37 of FIGURE 1 with, for example, a 50 ohm load on one output and an open circuit on the other, gives a 3:1 VSWR result, or in a ohm sum port Z impedance for a 50 ohm hybrid. In such a case, half of the power sent into the sum port is absorbed by the matched load, and the other half is distributed evenly between the difference port and the sum port, for isolations between output ports of about 20 db or greater. For ideally complementary loads, however, the same amount of power gets absorbed by each load, and all of the reflected power gets channelled into the difference port. We thus have 0.5 db complementarization loss for 2:1 load VSWR, 1.25 dbfor 3:1 VSWR, and so on.

These same power balances apply to the individual elements of an idealized complementary pair when fed by a 180' degree hybrid, and consequently to the total complementary pair element group. Obviously, the elements should be most nearly ideally complementary, i.e., have equal VSWR and be on opposite sides of the Smith Chart at all frequencies of the design band. The general impedance characteristics of such theoretical complementary pairs are depicted in FIGURE 16. The impedance locus of each element rotates around a certain center Z point in the complex impedance plane, which is preferably located on the real axis. The element characteristics are then approximately complementary with respect to this center point. When one element has a high impedance, the other one has a low impedance and vice versa. When such a pair of elements is fed with the same 180 degree hybrid, the sum port will be well matched over a very wide frequency range. Theoretically an infinite bandwidth would result, but in practice two factors limit the useful frequency range: The hybrid bandwidth and the E-plane pattern of the elements. The E-plane pattern, in particular, is only good up to about 0.6 wavelength monopole height, or 1.2 wavelengths dipole length. This is not really a limitation for the complementary pair as an element group in large, broadband phased arrays, particularly when beam steering to wide angles is required. In this case the array factor limits the over-all system bandwidth to about 4:1 frequency range anyway, because beam steering to, say, 40 degrees requires that the element spacing be less than 0.6x (wavelength) if grating lobes are to be avoided, since the spacing S has to be:

A 1+ in For a 4:1 frequency range, the spacing between elements is then 0.15 wavelength at the lowest frequency, and this should be the minimum tolerable spacing with respect to mutual coupling variation.

If broadband operation over a 4:1 frequency range is desired, the hybrid feeding the complementary pair has to have the same bandwidth. Such broadband hybrids are now available for a variety of frequency ranges and power levels. Particularly at HF and VHF, hybrids with up to :1 bandwidth and power levels up to 100 kw. average have already been built and tested. Using one of these high-power HF hybrids, for instance, a complementary pair element group can be designed giving performance approaching ideally a VSWR of 1.1:1 from, say, 6 to 24 megacycles, without the necessity of tuning the elements. The maximum insertion loss will be determined mostly by the (equal) VSWR of the individual elements because such high-power hybrids have insertion losses of typically less than 0.2 db. Assuming a 2:1 VSWR, which is constant with frequency, so that the element impedances rotate around a 50-ohm Smith Chart center, we get an impedance matching insertion loss of about 0.5 db when one of the elements is, e.g., ohms resistive and the other is 100 ohms resistive. The inequality of power split (in the transmitting case) is zero for this case, and very small for most other frequencies for isolations between the two output ports of 20 db or more. FIGURE 17 shows the measured values of insertion loss for a 20 kw. HF hybrid T, and FIGURE 18 the VSWR at the sum port of such a hybrid, when the other ports are matched or terminated in complementary impedances.

An example of naturally complementary elements are the dipole 41 and folded dipole 42 such as shown in FIGURE 19 mounted on a reflector 31 and above a ground plane 38 substantially the same as the similarly numbered reflector and ground plane in the FIGURE 3 embodiment. At very low frequencies, these elements are complementary, one approaching a short, the other an open circuit. In the resonant region between the first three resonances the two elements can be designed to be complementary at a certain frequency, and will stay fairly complementary over about an octave. Larger bandwidths are hard to achieve because of different rates of rotation on the Smith Chart, unless the VSWR specification at the hybrid sum port is somewhat relaxed. Thin dipoles and folded dipoles with a length-to-diameter ratio of about 75:1 are approximately complementary with respect to ohms, so that a 4:1 impedance transforming balun has to be employed before the two impedances may be combined in a 50 ohm hybrid.

It should be noted that in the broadside vertical dipole and folded dipole embodiment of FIGURE 19 giving a unidirectional group pattern, the elements are staggered vertically to minimize any existing detrimental effects of mutual coupling. An approximate optimum trade-off between overall complementary pair element group height and impedance match is provided when the top element is so positioned that its feed point is approximately the same height as the top of the lower element. Further, the folded dipole 42 is provided with an extended element top 43 for added top loading to provide fine adjustment of its impedance with respect to the dipole 41 to provide optimum complementarity over a 2:1 frequency range. The dipole 41 is provided with a feedline 44 connection through the reflector 31 and dipole balun box 45 which is fastened to reflector 3-1 by conventional means, and folded dipole 42 in like manner, is provided with a feedline 46 connection through the reflector 31 and folded dipole balun box 47, which is also fastened to the reflector 31 by conventional means. While individual element impedances of the FIGURE 19 embodiment have as high as 10:1 VSWR at the low end of the design band, if a sum port VSWR of about 3:1 can be tolerated, the frequency range is increased to over 4:1. The azimuth pattern of this complementary pair element group is that of a closely spaced pair in front of a reflector, and changes little with frequency. The general shape of this pattern resembles that of the endfire complementary pair element group, but is somewhat narrower, so that beam steering is only possible to about :30 degrees if this group is used in an electronically steerable array.

Please refer to FIGURES 20' and 21 for large arrays using complementary pairs with optimum properties with respect to installation ease and also array factors providing a desired operational bandwidth. FIGURE 20 shows a vertically polarized array, of interlaced monopoles 48 and folded dipoles 49, embodiment 50. The monopoles which are individually mounted on appropriate mounting supports '51 extend vertically with respect to their mountings from the ground plane 52. The monopoles 48, fed at the apex bottoms, are positioned in substantially the same plane as the folded dipoles 49 which are mounted in a common plane on a mount structure 53 of conventional nature. The folded dipoles 49 are fed at the cross points 54, detail not shown. Further, it should be noted that the monopole 48 and the folded dipole 49 elements are staggered vertically to minimize any existing detrimental effects of mutual coupling as with the embodiment of FIGURE 19. Further, the elements in a substantially common plane are spaced approximately 0.1x, of frequency bandwidth center frequency, from a reflector screen 55 of relatively large conventional construction mounted to extend between vertical towers 56 and 5-7. At high frequency, with a proper ground radio system excellent low angle coverage is obtained with this antenna over a 5:1 frequency range.

The embodiment '58 of FIGURE 21 is a horizontally polarized broadside reflector array which also has staggered elements to provide maximum bandwidth without grating lobe formation. This embodiment allows close element spacing and uniform reflector illumination and provides approximately a 4:1 frequency range. In this embodiment the dipoles 59 and folded dipoles 60 are mounted on a common supporting structure 61 in substantially a common plane which is spaced approximately 0.1x from reflector screen 62 mounted between vertical towers 63 and 64 mounted to extend vertically from the ground plane. Both the dipoles and folded dipoles of FIGURE 21 are fed at center element cross points 65 and 66, respectively, detail not shown. With both embodiments 50 and 58, the reflectors 55 and 62 result in a high front-to-back ratio, and a very low forward side lobe factor may be attained. These basic array module approaches may be extended to form broadside arrays of very high gain with the limit at high frequency generally being set by minimum tolerable beam widths.

If a reflector is not desirable, and low side lobes are not mandatory, then the endfire array of FIGURE 22 is a good antenna solution, although in this case the bandwidth is limited by the array factor to about an octave. With this embodiment, some gain optimization in side lobe control is possible with non-uniform spacing of complementary pair element groups. In the particular showing, three element groups '67 are of one longitudinal spacing over ground plane 68, with the end element groups 69, while of the same construction as element groups '67, being more widely spaced along the same longitudinal line outwardly from the closest element group 67 than the spacing between element groups 67. Because of the great extent in depth of this array, and resulting low effective elevation aperture, good low angle coverage is also provided. In each element group of this antenna array, a dipole 70 is mounted directly above a leg 71 of a folded monopole 72 by structural means not detailed. The dipole 70 of each complementary pair element group 67 and 69 is fed by a feedline 73 extended vertically through and insulated from the folded monopole leg 71 and through a portion of each dipole 70 to a center cross feed point 74'. The other legs 75 of each folded monopole 72 are fed at a lower conical apex feed end 76 at the ground plane '68. It should be noted that in order for the two elements in each group to come down to the 50 ohm level of a 180 degree feed hybrid, a different impedance transformation network is required for each element having a transformation ratio of approximately 211 for the dipole 70 and 4:1 for the folded monopole 72. Here again it should be noted that the elements are staggered inherently with the vertically stacked dipole and monopole construction employed.

The paracylinder antenna 77 for high frequency or very high frequency usage is shown in FIGURE 23 to employ an endfire monopole complementary pair element group 78 feed utilizing two spaced monopoles 79 and 80 fed through a 180 degree hybrid to their bottom conical apex feed ends 81 located at the ground plane 82. The feed beam direction is from the monopole elements 79 and 80 toward the paracylinder parabolic arc reflector 83. Reflector 83 is a relatively large structure provided with increasingly large spacing between wires 84 with increasing distance from the center of the paracylinder located approximately at vertical center reflector supporting tower 85 in the particular paracylinder reflector shown. Other towers 86A, 85B, 85C, and 85D are provided in increasing distances to each side from the reflector center mounting tower 85. The increasingly larger wire spacings between Wires 84 with increasing distance from the center tower 85 provides for controlled reflector 83 leakage with the outer portions of the reflector 83 becoming significantly effective only at the lower frequencies. The endfire monopoles 79 and 80 may be provided with guy wire installation support, not shown, in a conventional manner as required. This antenna feed system advantageously provides good illumination of the wide angle aperture while still having sufficient directivity to greatly minimize 12 gain variations due to feed mage interference. Furthermore, directive gains up to about 30 db are feasible at the high end of approximately a 4:1 operational bandwidth.

In the paraboloidal reflector antenna system embodiment 86 of FIGURE 24, a paraboloidal reflector 87 is provided with a dual complementary pair element group feed 88 consisting of two cross polarized endfire dipole complementary pair element groups to thereby be a system capable of providing all types of polarization. An advantage of this feed system over a log periodic structure in this embodiment is an almost constant phase center position through a wide frequency range of operation. It is an embodiment with a feed front-to-back ratio that can be improved by the addition of a small complementary pair element group reflector at the outer end of tube 89 with some increase in aperture blockage. The antenna 86 can provide very high gains at VHF and UHF with the upper limit being dictated by structural mechanical considerations. Depending upon the types of baluns used in and feedlines through tube 89, and hybrids employed in the feed circuit, approximately a 5:1 bandwidth range is achievable in frequency ranging up to approximately 1 gc. (gigacycle). For S-band and C-band applications octave frequency range would be feasible as limited primarily by balum circuit bandwidth limitations. Feed of this antenna is with lines coming through the back of the reflector 87 through tube 89 to balum circuitry in the vicinity of the dual complementary pair element group feed 88 at the outer end of the tube 89.

In FIGURE 25 a low-silhouette broadband fixed beam circular antenna array embodiment 90 is portrayed. This embodiment employs a plurality of endfire monopole complementary pair element groups 91 with each element group having a post-type reflector 92 associated therewith and in beam directional alignment with the monopoles 93 and 94 of the respective element groups 91. All the element groups 91 are oriented in substantially the same beam directional alignment and are arranged in, generally, a circular array of a multiplicity of element groups above ground plane 95. While this post-type reflector equipped multiple complementary pair group antenna array does not provide as much group directivity as a large flat reflector embodiment, the array factor does have a reasonably good front-to-back ratio. It is an antenna array embodiment having a large effective aperture both in azimuth and elevation and also good elevation beam steering potential. Azimuth steering with this embodiment is also possible through a limited angular range as limited by group pattern limitations. It is an antenna imposing large area requirements that may be justified when low antenna array height is desired along with ease feeding of the end-fire monopoles through the ground plane 95.

The electrically steerable complementary pair element group Wullenweber type antenna embodiment 96 of FIG- URE 26 is an example of a relatively simple antenna. structure capable of being steered electromechanically or electrically through an arbitrary range in azimuth, dependent upon the percentage of a full circle through which the reflector 97 and endfire complementary element groups 98 are provided. Each endfire complementary element group 98 includes endfire monopoles 99 and 100 in longitudinally spaced relation on radials and spaced outwardly from circular reflector 97 with the radials being radials relative to substantially the geographic center of the circular reflector 97. This antenna approach is most economically efficient for a full circle embodiment such as shown particularly when a full 360 degrees of azimuth steering electronic feeding control is required, such as for high frequency direction finding. Here, again, this embodiment employs endfire monopoles advantageously fed through or at the ground plane 101. The antenna reflector 97 is shown to be a multi-tower 102 supported screen or depending wire type reflector. It should be noted that horizontally polarized complementary pair element groups could be used in a Wullenweber antenna system in place of the endfire monoploe complementary element groups shown in the antenna embodiment 96 of FIGURE 26. Further, the bandwidth of the monopole array can be extended by vertically stacked low frequency elements. An antenna system is thereby provided wherein the entire high frequency spectrum may be covered in one structure with, however, no elevation steering possible. A further notable advantage of such complementary pair element group equipped Wullenweber-type antennas is that low minor lobe levels are achieved.

The circular antenna array embodiment 103 of FIG- URE 27 is equipped with endfire monopole complementary pair element groups 104 aligned on radials emanating from substantially the same image point. It has inner endfire monopoles 105 substantially on a circle, the outer endfire monopole elements 106 On a circle concentric with the circle of the inner monopoles 105, and with the two concentric point rings spaced a distance equal to substantially a quarter of a wavelength of the operational bandwidth center frequency. Here, again, in this embodiment the endfire monopoles are advantageously fed at or through the ground plane 107. This embodiment has improved operational side lobe conditions and allow 360 degrees azimuth steering over approximately a 3:1 frequency range. A further modification, not shown, is an extension of the circular array approach of FIGURE 27 to an additional dimension in expanding an array to a spherical array with hemispherical operational coverage.

If azimuth steering is required to approximately :50 degrees, the broadside reflector 108 endfire complementary pair element group antenna array embodiment 109 of FIGURE 28 is very eflicient structurally and is operationally an economical antenna. This employs a multiplicity of endfire monopole complementary pair element groups with one endfire element 110 of each group positioned substantially along a line approximately 0.1x of the operational bandwidth low frequency from the reflector 108 and parallel thereto. The other elements 111 of the element groups are in longitudinal alignment along a line substantially parallel to the line of the endfire elements 110 and substantially parallel to the reflector 108 and spaced an additional distance outwardly relative to the reflector 108 from the inner elements 110 approximately equal to 0.1251 low and 0.375) high frequency of the operational frequency bandwidth range. Furthermore, the element groups are spaced from one another in the range of from 0.2x low to 0.61 high frequency of the operational frequency bandwidth. Here, again, the endfire monopoles are advantageously fed at or through the ground plane 112, detail not shown.

Another interesting complementary pair element group hybrid system feed antenna embodiment 113 is shown in FIGURE 29 to be a self-supporting conductive material cylinder 114, miniature slot 115 and dipole 116 complementary pair element group antenna. The enclosed cavity chamber 117 of generally arcuate shape enclosed by conductive material defining such a cavity encloses dielectrical material fill 118 of ceramic or a plastic dielectric material completely filling the cavity with the exception of the coaxial inner stub feed 119 from coaxial line 120. The outer coaxial sheath of coaxial line 120 is connected to the conductive material forming chamber 117. The other coaxial line 121 passes through the interior of the antenna cylinder 114 to extend through a cylinder wall port 122 and interiorl upward through the lower section 123 of dipole 116 to a connection of the outer sheath of coaxial line 121 with the upper conical apex feed end of the lower section 123. The inner line of coaxial cable 121 is connected to the lower conical apex feed point of the upper section 124 of dipole 116.

The vertical dipole 116 in this embodiment is fed 90 degrees out of phase with slot 115, a feed phasing which may be accomplished with a 180 degree hybrid, such as hybrid 37 of FIGURE 1 and FIGURE 4 and other figures in the case, as part of a feed network including additional phasing circuitry. The additional phasing circuitry could be provided such as by making one of the coaxial lines and 121 a quarter of a wavelength longer than the other, as had been accomplished with the coaxial lines 35 and 36 with reference to FIGURE 4 and the embodiment of FIGURE 3. Dipole 116 may be mounted on cylinder 114 as by bracket 125 of non-conductive material in a conventional manner. In this antenna structure, the slot is made vertically physically shorter than half a wavelength, a factor eliminating any danger of elevation pattern lobing. It is an antenna useful for circular polarized UHF transmission that may employ adjustable position cover plates and inductive loading holes, detail not shown, at opposite ends of slot 115 for tuning as may be desired. While one element complementary group with the hybrid feeding system of this embodiment is shown on the cylinder 114, a multiplicity of such complementary pair element groups could be arranged around the circumference of cylinder 114 to provide desired azimuth coverage and, further, several of the circularly polarized groups may be stacked vertically to increase gain. While practically any reasonably desired level of gain and azimuth coverage may be thereby attained, such increase in the numbers of the complementary groups particularly with the stacking thereof is accomplished at some expense in efliciency in the individual element groups.

Thus, it may be seen that this invention provides very effective and efiicient antenna systems utilizing various degree hybrid feed complementary pair element group embodiments useful through extended frequency bandwidth in high frequency applications. Various high frequency antenna embodiments set forth would be particularly useful in fixed and mobile communications systems, in direction finding antenna usuage such as for air traffic control systems, and high quality broadcasting relay lengths, and in over-the-horizon radar installations. Furthermore, there are many uses to be found with various embodiments spanning VHF and UHF frequency ranges and to even higher frequency ranges. Further, various embodiments presented, with a constant phase center and constant radiated power with varying efiiciency, allow for transmission of narrow pulses with a minimum of dispersion. Various other applications include space tracking, spectrum surveillance and radio astronomy.

The complementary pair element groups in various antenna systems, particularly in the broadside and endfire versions are valuable new broadband radiators for both fixed beam and steerable beam antennas as major families. The fixed beam antenna family may be sub-divided further into focusing reflector equipped antennas such as paracylinders and paraboloids, circular arrays, broadside reflector arrays, and endfire arrays. The electically steerable antennas may be divided as indicated by various illustrated and suggested embodiments, for example, into circular and spherical arrays and into linear and planar arrays. Thus, a systematic classification is evolved placing the various antenna configurations of both the major families (fixed beam and steerable beam antennas) into various categories, depending on the type of complementary pair element group embodiment used,' and whether or not a reflector is embodied and so on, and would therefore appear to be a major advance in the state of the art of broadband antennas. Further, this is all accomplished with complementary pair element group element spacing, in the various embodiments, equal to less than approximately 0.7x of the operational bandwidth center frequency. This is a spacing limitation necessarily imposed to effectively achieve mutual radiation coupling required between paired elements in providing the complementary pair element groups.

Whereas, this invention is here illustrated and described with respect to specific embodiments thereof, it should be realized that various changes may be made without departing from the essential contribution to the art made by the teachings hereof.

I claim:

1. In an antenna system, a complementary pair element group having complementary impedance characteristics with element spaced less than approximately 0.7 wavelength of the operational bandwidth center frequency so as to be effectively radiation mutually coupled; a feed network between each of the elements of the complementary pair of elements; and with the feed network including a Magic T hybrid junction having a difference port terminated in an impedance substantially equal to the characteristic impedance of the hybrid difference port and having two additional ports connected to respective elements of the complementary pair of elements, with the two connecting cables matched to the two additional ports, whereby the reflection coefficients cancel.

2. The complementary pair element group of claim 1, wherein the Magic T hybrid junction is a 180 degree hybrid junction; and the two elements of the complementary pair element group are additive in radiation characteristics primarily in one direction.

3. The complementary pair element group of claim 2, wherein the group is an endfire complementary pair element group mounted above a ground plane; and means feeding the two elements of the endfire complementary pair in the vicinity of the ground plane.

4. The complementary pair element group of claim 3, wherein an element of the pair of elements is provided with a top loading extended planar end.

5. The complementary pair element group of claim 3, wherein the two additional ports of the 180 degree hybrid junction are connected to the respective elements through coaxial lines; and with one of the two coaxial lines substantially a quarter of a wavelength of the operational bandwidth center frequency longer than the other coaxial line.

6. The endfire complementary pair element group of claim 5, wherein a planar reflector is provided perpendicular to the longitudinal axis plane of the endfire paired elements; with said planar reflector also substantially parallel to the axis of each of said endfire paired elements; and with the reflector having a smaller spacing from the closest element than the spacing between the paired elements.

7. The complementary pair element group of claim 2, wherein the complementary pair element group includes, a vertical dipole and folded dipole mounted in spaced relation from a planar reflector element and above a ground plane.

8. The vertical dipole and folded dipole complementary pair element group of claim 7, wherein the feed connections are through impedance transforming baluns at the mounting of each respective element; an extended element top is provided on the folded dipole; and the elements are staggered vertically to minimize any existing detrimental eflects of mutual coupling.

9. The complementary pair element group of claim 2 wherein a plurality of complementary pair element groups is used in a vertically polarized antenna array including interlaced monopoles and folded dipoles; with the monopoles mounted to extend vertically from a ground plane in substantially a common plane; the folded dipoles being mounted in complementary pair element group relation with the respective monopoles in substantially the common plane of the monopoles; and a reflector screen spaced approximately 0.1 wavelength of the operational bandwidth low frequency from the common plane of the monopoles and folded dipoles.

10. The complementary pair element group of claim 2 wherein a plurality of complementary pair element groups is used in a horizontally polarized antenna array including interlaced monopoles and folded dipoles; with the monopoles and the folded dipoles being mounted in a 16 complementary pair element group relation in substantially a common plane; and a reflector screen spaced approximately 0.1 wavelength of the operational bandwith low frequency from the common plane of the monopoles and folded dipoles.

11. The complementary pair element group of claim 2, used in plurality in an endfire array as a multiplicity of complementary pair element groups; with, in each complementary pair element group, a folded monopole mounted to extend above a ground plane common of the whole antenna array;a dipole mounted above a leg of the folded monopole of that complementary pair; and with a feedline extended vertically through a folded monopole leg to the dipole.

12. The complementary pair element group endfire array of claim 11, wherein the complementary pair element groups are in substantially aligned relation with a plurality of groups having one spacing, and with terminating end groups having a greater spacing from adjacent complementary pair element groups.

13. The complementary pair element group of claim 2, wherein an endfire monopole complementary pair element group feed is provided for a paracylinder reflector equipped antenna with the complementary pair element group feeding toward the reflector.

'14. The paracylinder antenna of claim 13, wherein the reflector is a parabolic arc reflector built up of reflective elements; with increasingly large spacing between reflective elements of the parabolic reflector with increasing distance to each side of the center of the parabolic arc reflector so that outer portions of the reflector become significantly and increasingly efl'ective only at the lower frequencies.

15. The complementary pair element group of claim 2, wherein dual complementary pair element groups are constructed as two cross polarized endfire dipole complementary pair element groups in a feed for a paraboloidal reflector antenna.

16. The complementary pair element group of claim 2, used in plurality in the form of endfire monopole complementary pair element groups in a low silhouette broadbeam-fixed-beam generally circular antenna array; with each element group being provided with a post-type reflector positioned in operational association therewith and in beam directional backing alignment with the respective endfire monopole complementary pair element groups.

17. The complementary pair element group of claim 2, also including a Wullenweber-type antenna reflector presenting an effectively circularly arcuate electronic radiation reflective surface; and with endfire monopole complementary pair element groups positioned substantially on radials of the circularly arcuate reflector image center and outwardly from the reflector.

18. The complementary pair element group of claim 2, used in plurality in the form of endfire monopole complementary pair element groups in a circular antenna array with the inner endfire monopoles of the multiplicity of element groups on a common circle; and the outer endfire monopole elements on a circle concentric with the circle of the inner monopoles; and with the two concentric circles having a spacing substantially equal to a quarter of a wavelength of the operational bandwidth frequency; and with the endfire monopoles fed substantially at the ground plane of the antenna array.

19. The complementary pair element group of claim 2, wherein a broadside endfire complementary pair element group antenna array employs a multiplicity of endfire monopole complementary pair element groups with one element of each group common to a line approximately one-tenth of a wavelength of a low frequency of the operational frequency bandwidth range from a reflector; and the other elements of the element groups substantially common to a second line also parallel to the reflector and spaced a predetermined additional distance outwardly relative to the reflector from the inner elements approximately equal to 0.125 wavelength of a bandwidth range 17 18 low frequency and to 0.375 wavelength of an operational OTHER REFERENCES frequency bandwidth high frequency.

20. The complementary pair element group of claim New High-Frequency Antenna. The Passive Network 2, wherein a miniature slot and dipole complementary Y Dunlavy Electronics, 1964, pair element group is used in an antenna cylinder con- 5 pp. 34-36.

tained slot and dipole complementary pair element group antenna system; with the dipole mounted outwardly from ELI LIEBERMAN, Primaly Examiner.

the cylinder, and displaced to one side from the slot open- MARVIN NUSSBAUM, Assistant Examiner.

ing in the cylinder Wall.

.8. C1. References Cited 10 343 794 840 844 8 1 8 854 5:6 UNITED STATES PATENTS 2,198,604 4/1940 Everitt 343816 UNITED STATES PATENT OFFICE CERTIFICATE OF CORRECTION Patent No. 3,449,751 June 10, 196! Klaus G. Schroeder It is certified that error appears in the above identified patent and that said Letters Patent are hereby corrected as shown below:

Column 1, line 17, "imepance" should read impedance Column 3 line 11 "reactant" should read reactance Columr 4, Equation (1), should appear as shown below:

2 2 2 N ng .l 4. +EJ.A

is Z Z Z Z 2 2 N +N l (l l) 2 2*"; T 2

Column 4 Equation (2) should appear as shown below:

1 1 2 Z Z Z Z 2 l l l 1 Z Z l l Z Z Z Column 5, line 63, after "forward" insert element Column 6 llne 9 "is" should read it line 14 "that" should read the line 61 "This" should read Thus line 70 "approximately" should read approximating Column 7 Equatic (1A) should appear as shown below 1 Z 2 (N (N +N Y Y N 2 (Y +Y Y Y 1 1 l l A l l A 2 2 2 1: 2 (N N Y (N Y N Y Column 12 line 1 "mage" should read image Column 13 

